1. Field of the Invention
The present invention relates to electrical motor control systems and, more specifically, to a circuit that measures torque and flux current in a synchronous motor.
2. Description of the Prior Art
Electric motor drive applications are increasingly used in automotive applications. Recent trends include the use of electric motors in power steering. Since the battery voltage standard is only 12v, the peak motor power is limited to a low horsepower rating, typically on the order of a 1-2 HP motor. In high performance motion control, such as electric power steering, low torque ripple, low cost, small size and high reliability are required. These factors typically lead to the use of a permanent magnet synchronous motor (PMSM) instead of DC motor, a switch reluctance motor or an induction motor. The magnetic field and magneto-motive forces (MMF) in the PMSM are assumed to be sinusoidally distributed in space in order to minimize torque ripple.
In a PMSM drive, motor torque feedback is required to have a precise torque control. A vector control is typically used to achieve a high performance motor drive system. Speed sensor feedback is required and is easily obtained, for example, through a motor shaft encoder at relatively low cost. However, to control motor torque precisely, an absolute rotor position sensor is also required, which is obtained from sensing a motor back electro-motive force (EMF) or from a resolver. A motor torque feedback signal can be obtained from a torque sensor, but including a torque sensor may be a costly solution. A more commonly used method is to sense the motor currents and then derive the motor torque feedback from the sensed currents. However, existing methods of sensing motor currents are difficult, can be costly and can reduce performance.
The steady state electromagnetic torque (Te) of PMSM, expressed in the dq model, is:       T    e    =            3      2        ⁢          P      2        ⁢                  1                  ω          e                    ⁡              [                                            E              1                        ⁢                          I              qd                                +                                    (                                                X                  dx                                -                                  X                  qs                                            )                        ⁢                          I              ds                        ⁢                          I              qs                                      ]            
Since there is no damper winding in PMSM, the torque equation is also valid for the instantaneous case.
In an air gap magnet motor, where Xds≈Xqs, the torque expression is reduced to:       T    e    ≈            3      2        ⁢          P      2        ⁢          1              ω        e              ⁢          E      I        ⁢          I      qs      
The internal back EMF peak voltage EI (volt) is given as:
EI=Kvxcfx89e
And the quadrature axis (q-axis) motor current Iqs (amp) is:
Iqs=Iqds cos xcex3
Hence, the torque can also be expressed as:
Te≈KTIqds cos xcex3
Where,
P=Number of poles.
xcfx89e=Electrical synchronous motor speed which is also the stator frequency in rad/sec.
Xds=Direct axis reactance encountered by the d-axis and q-axis current components in Henry-rad/sec.
Xqs=Quadrature axis reactance encountered by the d-axis and q-axis current components in Henry-rad/sec.
Xs=Xds=Stator referred synchronous reactance in Henry-rad/sec.       r    s    =                              Short          ⁢                      xe2x80x83                    ⁢          Circuit          ⁢                      xe2x80x83                    ⁢          Load          ⁢                      xe2x80x83                    ⁢          Loss                ⁢                  xe2x80x83                                      (                      Short            ⁢                          xe2x80x83                        ⁢            Circuit            ⁢                          xe2x80x83                        ⁢            Armature            ⁢                          xe2x80x83                        ⁢            Current                    )                2              =          effective      ⁢              xe2x80x83            ⁢      stator      ⁢              xe2x80x83            ⁢              res        .            
Kv=EMF constant in V/(rad/sec.).
KT=Torque constant in Nm/Amp.
Iqds=Amplitude motor stator peak current in Amp, hence it is a dc quantity.
Vqds=Amplitude motor stator peak phase voltage in volt, hence it is a dc quantity.
xcex3=A space angle measured at the vector position of the current Iqds with respect to the q-axis (where EI is located). Cos (xcex3) is defined as internal power factor. Angle xcex3 is positive if the current vector Iqds leads the voltage vector EI. It is also referred as torque angle.
xcex4=A space angle measured at the vector position of the Vqds with respect to the q-axis (where EI is located). It is sometime called as xe2x80x9cphase advancexe2x80x9d. Angle xcex4 is positive if the voltage vector Vqds leads the voltage vector EI.
xcfx86=A space angle measured at the vector position of the Iqds with respect to the Vector position of the current Vqds. Cos (xcfx86) is defined as load power factor. Angle xcfx86 is positive if the current vector Iqds leads the voltage vector Vqds.
FIG. 1 shows a typical space vector representation 10 of 3-phase PMSM operation. Positive values for angles xcex3, xcex4 or xcfx86 mean the angle is oriented counter clockwise with respect to the corresponding reference q and d axis. Iqds, Vqds, Ei vectors along with the corresponding reference q and d axis, are simultaneously moving in a counter clockwise direction when the speed is positive. This means that the vectors rotate one 360xc2x0 turn when the motor rotates one electrical turn. In the time domain expression, the phase sequence is A-B-C which corresponds to positive speed rotation. In a positive speed motoring operation, a PMSM typically operates where current vector Iqds is in the first quadrant, i.e., xcex3=0 up to base speed operation and 0xc2x0 less than xcex3 less than 90 beyond base speed operation.
To summarize, the two requirements for vector control in PMSM are measurement of rotor position (absolute rotor position is required) and precise control of the stator current to correctly position the resultant stator MMF in relation to the rotor position. Therefore, it is important to control the magnitude Iqds and the angle xcex3 independently. The torque response will follow the stator current Iqds instantaneously so long as the angle xcex3=0xc2x0. If the angle xcex3 attains a value other than zero, there will be a component of the stator current in the field axis (d-axis) and a flux change will take place. Since the flux change is not instantaneous, the torque response will also not be instantaneous if angle xcex3 or Iqds is changed.
One way to control amplitude and phase of stator current independently is to use a current regulated PWM inverter (CRPWM) in a stationary reference frame. The CRPWM provides a conceptually simple means for implementing torque control with independent q-axis and d-axis current inputs (Iqs and Ids). In essence, all that is required is to use absolute rotor position information to convert the Iqs* and Ids* commands in the rotor reference frame to a stator reference. The stator referred currents, at stator frequency, become the current commands for CRPWM. However, this technique requires instant stator current feedback information obtained through current sensors. In addition, the bandwidth of this regulator must be relatively high including dc in case of a stall mode condition.
Another method to measure q-axis and d-axis current inputs (Iqs and Ids) independently is by using a current regulator in rotor reference frame, i.e., a synchronous reference frame, since the rotor frequency is the same as the stator frequency. In this method, the Iqs* and Ids* commands are dc quantities. Such regulators typically do not require a relatively high bandwidth. Iqs and Ids feedback are required and in steady state they are dc quantities. Iqs and Ids feedback are typically derived from stator phase currents feedback, ias, ibs, ics, using Park""s transformation.
In small horsepower drives, current sensors such as Hall Effect devices or current shunts are often placed in series with motor phases. In a three-phase system, two of such sensors are required. Such devices introduce a significant cost relative to the system cost. A lower cost method is to use a current shunt placed in the dc link.
In a closed-loop current mode motor control, one method is to sample dc link current idclink and, knowing the corresponding PWM inverter switching state, to decode the stator phase currents, ias, ibs, ics. With the rotor position information, the instantaneous motor torque current Iqs and motor flux current Ids can be derived from the stator phase currents. However, this method requires a significant amount of processing and sampling idclink at a high sample rate, typically at the PWM switching frequency, at about 20 kHz. Therefore, a simple micro controller (MCU) implementation is impossible with this method. A high performance MCU or digital signal processor (DSP) with fast analog-to-digital (A/D) converters is required to accomplish the task.
A closed-loop current mode motor control, such as previously described, requires motor torque and flux currents obtained at high sample rate. In applications where motor torque and flux current feedback are only used for safety check diagnostic, computation integrity check, or feed forward control algorithm, Iqs and Ids are not required to be acquired at high sample rate. Therefore, there is a need for a simpler method for cost effective implementation.
In a three-phase system, in one electrical cycle, there are six positions where the back EMF occurs at the maximum or minimum value. If the motor stator currents are acquired at the instant where the back EMF is at peak (maximum or minimum), theoretically the torque current Iqs can also be obtained. In one electrical cycle, in theory six torque current Iqs measurements can be obtained. If the motor stator currents are acquired at the instant where the back EMF is at 90xc2x0 beyond its peak, in theory six flux current Ids measurements can also be obtained in one electrical cycle. Six Iqs and 6 Ids feedback measurements are sufficient in applications where motor torque and flux currents feedback are used only for a safety check diagnostic, a computation integrity check, or a feed forward control algorithm.
One existing method uses the peak detect method. Motor torque current Iqs is obtained by sampling the peak value of dc link current idclink at the instant where the internal back EMF EI is at the peak (maximum or minimum). However the Iqs measurement is only accurate for a limited range of torque angle xcex3, xe2x88x9230xc2x0 less than xcex3 less than 30xc2x0, and phase advance angle xcex4, xe2x88x9260xc2x0 less than xcex4 less than 60xc2x0.
Unfortunately, torque angle xcex3 needs to be increased above the 30xc2x0 limit when rotor speed goes up. In PMSM, maximum rotor speed is obtained when xcex3 approaches 90xc2x0. The requirement on phase advance angle xcex4 is not a big constraint because it can be kept under 60xc2x0 in machine design.
In addition, the peak detect method will pose accuracy issues if there is an overshoot current in dc link current idclink due to diode reverse recovery current
Another disadvantage of this sampling technique is that it is not capable of measuring the d-axis current Ids directly. Ids can only be obtained by calculation from measured Iqs and command angle xcex3. Therefore, Iqs will not be measured accurately.
Therefore, there is a need for a cost effective method that measures torque and flux current accurately on a wider range of xcex3 angle without dc link current high sample rate requirement.